Microphone preamplifier

ABSTRACT

A microphone preamplifier, comprising a differential input ( 102 ) stage with a first and a second input terminal and an output stage with an output terminal; where the microphone preamplifier is integrated on a semiconductor substrate. A feedback circuit, with a low-pass frequency transfer function ( 103 ), is coupled between the output terminal and the first input terminal and integrated on the semiconductor substrate. The second input terminal provides an input for a microphone signal ( 105 ). Thereby a very compact (with respect to consumed area of the semiconductor substrate), low noise preamplifier is provided.

CROSS-REFERENCE TO PRIOR APPLICATION

The above-referenced application is the U.S. National Phase ofinternational Patent Application PCT/DK2004/000707, filed Oct. 14, 2004,which claims priority from U.S. Provisional Application No. 60/510,145,filed Oct. 14, 2003, and U.S. Provisional Application No. 60/542,305,filed Feb. 9, 2004, which is incorporated by reference herein. TheInternational application was published in English on Apr. 28, 2005 asWO 2005/039401 A1.

INTRODUCTION

The preferred type of microphone for telecom applications (ie Mobilephones) has for many years been electret microphones. This type ofmicrophone is based on the principle of a capacitor, which is formed bya movable member that constitutes a membrane of the microphone andanother member, eg a so-called back plate of the microphone. One of themembers of the microphone, preferably the membrane, is charged by aconstant electrical charge.

A sound pressure detected by the microphone will cause the membrane tomove and consequently change the capacitance of the capacitor formed bythe membrane member and the other member. When the charge on thecapacitor formed by these two members is kept constant, the voltageacross the two capacitor members will change with the incoming soundpressure level. As the charge on the microphone capacitor has to be keptconstant to maintain proportionality between sound pressure and voltageacross the capacitor members, it is important not to load the microphonecapacitance with any resistive load. A resistive load will discharge thecapacitor and thereby degrade or ruin the capacitors performance as amicrophone.

Therefore, in order to pick up a microphone signal from the capacitor,amplifiers configured with the primary objective of providing high inputresistance are preferred to buffer the capacitor from circuits which areoptimized for other objectives. The amplifier connected to pick up themicrophone signal is typically denoted a preamplifier or a bufferamplifier or simply a buffer. The preamplifier is typically connectedphysically very close to the capacitor—within a distance of very fewmillimeters or fractions of millimeters.

For small sized microphones only a very limited amount of electricalcharge can be stored on one of the microphone members. This furthers therequirement of high input resistance. Consequently, the input resistanceof preamplifiers for small sized microphones has to be extremely high—inthe magnitude of Giga ohms. Additionally, the input capacitance of thisamplifier has to be very small in order to achieve a fair sensitivity tosound pressure.

Traditionally, this buffer amplifier or preamplifier has beenimplemented as a simple JFET. The JFET solution has been sufficient, butdemands in the telecom industry call for ever smaller microphones—withincreased sensitivity. This yields a contradiction in terms sincesensitivity of the microphone capacitor drops as size goes down. Allother things being equal, this will further reduce the sensitivity ofthe microphone and the buffer in combination. The demands in the telecomindustry are among other things driven by market trends which encompasshands free operation of different types of small-sized equipment andmore widespread application of microphones in eg camera/videoapplications.

Today telecom microphones typically have a sensitivity of −40 dBV and acapacitance of 7 pF. This capacitance is also denoted the cartridgecapacitance. The term ‘cartridge’ is used to designate the microphonewithout the preamplifier. The vast majority of JFETs for this purposehave an input capacitance of approximately 5-7 pF. Comparing this inputcapacitance to the cartridge capacitance it can be deduced that half ofthe microphone signal is lost in being acquired by thepreamplifier—before it is being amplified.

Telecom microphones with an integrated preamplifier are sold in highvolumes and at very low prices. As cost of an amplifier for a telecommicrophone is directly related to the size of the preamplifier chip dieit is important, for the purpose of reducing price, that thepreamplifier die is as small as possible.

So obviously, there is a need for microphone preamplifiers with gain andvery low input capacitance, and lowest possible preamplifier die area.Additionally, low noise is important. Low noise is important as noisecan be traded for area—ie if the circuit has low noise and a noise lowerthan required, this noise level overhead can be traded for lower chipdie area and it is thus possible to manufacture the preamplifier atlower cost.

CMOS microphone preamplifiers have proven to be superior to JFETmicrophone preamplifiers in resent years since they can be designed withlower noise and lower input capacitance while providing a fair amount ofgain.

In hearing aids, microphone preamplifiers are exclusively designed inCMOS. The reason for this is that CMOS amplifiers designed in CMOStechnology provides signal to noise ratios far beyond what can bereached by amplifiers implemented in JFET technology. This applies inparticular when the cartridge capacitance is very low.

When designing a preamplifier in CMOS technology for a microphone thereis normally three noise sources. These sources are noise from a biasresistor, 1/f noise from an input transistor, and white noise from theinput transistor. We assume that input transistor noise dominates. Bothwhite noise and 1/f noise can be minimized by optimizing the length andthe width of the input transistor(s). This applies for any input stageeg a single transistor stage or a differential stage. The noise from thebias resistor can also be minimized. If the bias resistor is made verylarge then the noise from the resistor will be high pass filtered andthe in-band noise will be very low. This has the effect though that thelower bandwidth limit of the amplifier will be very low. This can be aproblem as the input of the amplifier will settle at a nominal valueonly after a very long period of time after power up. Additionally,signals with intensive low frequency content arising form eg slamming ofa door or infra sound in a car can overload the amplifier. Anotherrelated problem is small leakage currents originating from mounting ofthe die inside a microphone module. Such currents will due to theextreme input impedance establish a DC offset. This will reduce theoverload margin of the amplifier.

RELATED ART

Several solutions have been proposed using preamplifiers based on JFETor other technologies. However, these solutions do inherently have arelatively high noise level. Consequently, the prior art solutions cannot be designed for smallest possible die area.

In hearing aid microphones the problem of excessive sensitivity toinfrasound is handled by a two stage configuration with a bufferamplifier as input stage followed by a high-pass filter. When the signalhas been high-pass filtered it doesn't contain the large low frequencycomponents which could overload the amplifier and it can then beamplified further. This approach has proven to work well with hearingaid microphones where the cartridge sensitivity is relatively high egabout 20 mV/Pa.

For telecom microphones the cartridge sensitivity is normally about asless as 5-7 mV/Pa. However, new types of applications require amicrophone sensitivity of about 40 mV/Pa or more. Themicrophone/preamplifier configurations known from the field of hearingaids would—from a technical perspective—work fine, but for telecomapplications these configurations would be far too expensive as theyrequire a substantial amount of chip die area relative to the costdictated chip area available for telecom microphones.

The two stage configuration has two disadvantages; as it has two stages,it is noisier and because there is no gain in the first stage thephysical size of the high pass filter has to be relatively large. Itshould be noted that Noise and area are directly related. The size ofthe filter could be minimized by increasing the gain of the first stage,but the amplifier would be sensitive to overload because of lowfrequency components which are not diminished until in the subsequenthigh-pass filter. Thus the solution originally developed for hearing aidmicrophones will be far from optimal for the new high sensitive telecommicrophones. The area of the amplifier die would simply be too large andthe device consequently too costly.

The demand for smaller microphones and consequently the application ofsmaller microphones results in a smaller microphone capacitance. Thiswill in turn increase the spectral density of noise within the audiorange. Therefore a larger bias resistor is required to compensate forthe otherwise increased noise density in the audio range.

The demand for a large input bias resistor requires a small inputleakage current. Such a small input leakage current can only be obtainedby CMOS technology. In order to achieve a good signal-to-noise ratio,CMOS technology combined with a bias resistor larger than 10 GOhm ispreferred. In hearing aid applications the above is provided by a simple0-dB buffer in CMOS technology combined with a large bias resistor. Thiswill provide a feasible design since the sensitivity of microphones forhearing aid applications generally is relatively high.

However, since sensitivity is traded for low prise, microphones fortelecommunications purpose are less sensitive. From a marketperspective, there is a demand for a larger sensitivity of themicrophone and preamplifier in combination. So, therefore the gain inthe preamplifier is to be increased to meet the demand. Additionally,there is a demand for low noise in the audible range. Moreover, in orderto ensure a good signal-to-noise ratio while meeting the demand for arelatively large sensitivity, the input capacitance of the preamplifiermust be small to avoid an unnecessary signal loss from the microphone(cf. the equivalency of the microphone signal being exposed to a voltagedivider constituted by the capacitances).

Since, the chip area occupied by the preamplifier must be as small aspossible to obtain relatively low cost; the preamplifier must be assmall as it can be. Therefore, since amplifier configurations known fromhearing aids are generally not optimised for chip area to the sameextent, these configurations are not applicable. Further, one shouldbear in mind that buffers or amplifiers applied in hearing aids are notconfigured to provide such high gain levels as are required for thelow-sensitivity microphones used in telecommunication applications. Inhearing aids chips, more space is required for the same noiseperformance since buffers are required to avoid overload in hearingaids.

In the above, various aspects of known microphone preamplifierconfigurations have been discussed in the light of relevantsemiconductor technologies for implementing the configurations.

As the total sensitivity of the new types of telecom microphones arevery high, the output signal swing of the microphone can become verylarge; eg about 1 Vpp, which is far beyond what seen previously in bothtelecom and hearing aid market. The fact that the telecom microphone hasto be operated with two terminals (combined out and power) makes thedesign for a large maximum output signal swing even more difficult. Thiscalls for solutions different from hearing aid applications.

The intrinsic sensitivity of telecom ECM's are normally relatively low.As a consequence of this the preamplifier for a telecom ECM requiresgain.

Furthermore the telecom market today requires even higher sensitivitythan before. The consequence of this is that even higher gain from thepreamplifier is required. But still the same overload margin isrequired. Also the ability to handle large low frequency signals such ascar rumbleling and door slamming should be the same.

SUMMARY OF THE INVENTION

Thus, it is an objective of the present invention to provide apreamplifier with the lowest possible input capacitance, lowest possiblenoise, largest output signal swing in a two-terminal configuration andat the same time exhibiting the lowest smallest possible chip area.

It is an objective of the present invention to provide a preamplifierhaving a large power supply rejection and low distortion.

It is an objective of the present invention to provide an amplifierwhich is able to handle slowly varying signals with relatively largeamplitude at its input terminal while at the same time being able toamplify a low level signal with a higher frequency with low distortion.

It is an objective of the present invention to provide an amplifierwhich performance is very insensitive towards leakage and parasiticcouplings connected to the input.

The invention relates to a microphone preamplifier, comprising adifferential input stage with a first and a second input terminal and anoutput stage with an output terminal; where the microphone preamplifieris integrated on a semiconductor substrate; and a feedback circuit, witha low-pass frequency transfer function, coupled between the outputterminal and the first input terminal and integrated on thesemiconductor substrate. The second input terminal provides an input fora microphone signal.

Thereby a semiconductor microphone preamplifier is provided with afilter feedback configuration. This preamplifier can provide a largeloop-gain outside the audio band and will give rise to very littledistortion in the audio band. But more importantly, inter-modulationdistortion introduced by frequency components at low frequencies,outside the audio band, will be very low. The loop-gain characteristicprovided by the feed-back configuration provides e.g. lower distortion.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 shows a microphone comprising a preamplifier with a feedbackfilter;

FIG. 2 shows a source-follower coupled to a microphone and illustratesnoise sources in a microphone amplifier;

FIG. 3 shows the spectral noise density of the source-follower;

FIG. 4 shows plots of the transfer function for open loop amplifiergain, feedback filter gain, loop gain and preamplifier gain;

FIG. 5 a shows a first feed-back filter;

FIG. 5 b shows a second feed-back filter for IC implementation;

FIG. 5 c shows a fourth feed-back filter for IC implementation;

FIG. 5 d shows a third feed-back filter for IC implementation;

FIG. 6 shows a detailed view of an amplifier;

FIG. 7 a shows an amplifier with a feed-back filter and an inputclamping circuit;

FIG. 7 b shows a diode based input clamping circuit;

FIG. 7 c shows a PMOS based input clamping circuit;

FIG. 7 d shows a NPN transistor based input clamping circuit;

FIG. 8 a shows an amplifier with a feed-back filter and an output stage;

FIG. 8 b shows common source output stage;

FIG. 8 c shows source follower output stage;

FIG. 8 d shows an output stage with a combined common source and sourcefollower configuration;

FIG. 8 e shows a cascaded common source output stage;

FIG. 9 shows an amplifier configuration with RF filters;

FIG. 10 a shows an amplifier configuration with DC level compensationwithin the feed-back filter and within the input stage of the amplifier;

FIG. 10 b shows an amplifier configuration with DC level compensation atthe input terminal of the amplifier;

FIG. 10 c shows a circuit which implements a high-ohmic resistor;

FIG. 10 d shows an amplifier configuration with DC level compensation atthe input stage of the amplifier;

FIG. 11 shows an amplifier configuration with a voltage pump;

FIG. 12 shows a microphone comprising an electret microphone member anda differential amplifier with a feedback filter;

FIG. 13 a shows a differential amplifier with input and output terminalsand signals illustrating low frequency behaviour of the differentialamplifier;

FIG. 13 b shows the differential amplifier with input and outputterminals and signals illustrating a high frequency behaviour of thedifferential amplifier;

FIG. 14 shows a portion of a digital microphone comprising an electretmicrophone member and a differential amplifier in a first configuration;

FIG. 15 shows a portion of a digital microphone comprising an electretmicrophone member and a differential amplifier in a secondconfiguration;

FIG. 16 shows a portion of a digital microphone comprising an electretmicrophone member and a differential amplifier with a feedback filter;

FIG. 17 shows a preferred embodiment of a feedback filter;

FIG. 18 is a schematic view of a microphone with an integrated circuitand a microphone member; and

FIG. 19 is a schematic view of a microphone with an integrated circuitand a MEMS microphone member.

FIG. 1 shows a microphone comprising a preamplifier with a feedbackfilter. The microphone 101 comprises a microphone member Cmic, 105biased via a resistor Rb, 104 coupled to a voltage source 108 thatreceives a current input via the output terminal, designated Pwr/Out, ofthe microphone 101. In an alternative configuration, the resistor Rb,104 is connected to a ground reference and the microphone is biased byproviding a DC offset to an input terminal of the amplifier 102. Thus,the output terminal serves to provide a bias voltage Vb, via the voltagesource 108 and operating power to the microphone preamplifier 102 andits feedback filter 103 and to provide a microphone output signalresponsive to a sound pressure on the microphone member Cmic, 105.

A circuit node established at the interconnection of the microphoneCmic, 105 and the bias resistor Rb, 104 is connected to a non-invertinginput (+) of the operational amplifier 102. The amplifier 102 isprovided with a feedback circuit 103. The feedback circuit 103 has ininput port designated ‘a’ coupled to receive an output signal from theamplifier 102 and an output port designated ‘b’ coupled to an invertinginput (−) of the amplifier 102. The preamplifier comprising theamplifier 102 and feedback circuit 103 is implemented on a semiconductorsubstrate 107.

The amplifier 102 and the feedback circuit 103 have, in combination, afrequency transfer function from the non-inverting input (+) to theoutput (which corresponds to the circuit node connected to the inputport ‘a’ of the feedback circuit). This frequency transfer function hasa high-pass characteristic.

However, the feedback circuit has a frequency transfer function from theport ‘a’ to the port ‘b’ with a zero and a pole; wherein the zero islocated at a higher frequency than the pole. Thus the feedback circuithas a low-pass characteristic.

The feedback circuit in the form of a filter can be a first order filteror it can be of higher order; eg second order, third order or fourthorder. Also it can be implemented as a passive circuit or as an activecircuit. The feedback loop assures that the overall gain of theamplifier with feedback is relatively low at low frequencies andrelatively high at audio band frequencies.

The preamplifier is powered from its output terminal designated Pwr/Out.The amplifier is coupled as a non-inverting amplifier with themicrophone connected to the non-inverting input. This ensures that thecapacitive loading of the microphone is very low. Due to the feedback,the inverting input terminal (−) of the amplifier 102 will exactlyfollow the non-inverting terminal (+). If the input stage of theamplifier 120 is a differential transistor pair (i.e. a differentialstage), gate-source voltages of the transistor pair will remain constantand the input capacitance will consequently be very low. A more detaileddescription of one possible embodiment of an amplifier with adifferential input stage is provided in connection with FIG. 6.

To set the output DC level of the amplifier a DC offset can be buildinto the amplifier or even better into the filter. An implementation ofan amplifier with build in offset is explained in FIGS. 5 d, 10 a, 10 band 10 c.

FIG. 2 shows a source-follower coupled to a microphone cartridgeelement. The microphone cartridge element 201 is illustrated as acircuit model with a voltage generator 203 coupled in series with acapacitor Cmic, 204. The source-follower 202 comprises an active devicein the form of a PMOS device T, 206 which is coupled to a groundreference by means of its drain terminal and to a supply voltage Vdd viaa source resistor Rs, 207. The input of the source-follower is providedat the circuit node established by the gate terminal of the activedevice, the bias resistor, Rb, and the capacitor Cmic.

This circuit is a simple circuit it and is very useful for explainingthe basic function of a microphone preamplifier. This circuit has threesignificant noise sources; ie noise from the bias resistor Rb, whitenoise from the PMOS device T, 206 and 1/f noise from the PMOS device T,206.

The figure shows a very simple amplifier, but all amplifiers have thesethree significant noise sources.

FIG. 3 shows the spectral noise density of the source-follower. Thespectral noise density is shown as a function of frequency, f. The noisedensity is shown together with the so-called A-weighing curve.

A source follower has been used as an example but the spectral densityof the noise will have the same shape for any CMOS amplifier. Theguidelines for optimizing the spectral noise density for bestsignal-to-noise ratio, SNR, are basically also the same for other typesof amplifiers. It can be described how to minimize each component of thespectral noise density.

Firstly, the bias resistor generated noise component is explained. Onthe figure it can be seen where this noise component is dominating—seethe frequency range designated ‘Rbias’. The total noise power comingfrom the bias resistor can be calculated as kT/C where k is Bolzmansconstant, T is temperature in Kelvin and C is the capacitance connectedto the resistor; normally dominated by the microphone capacitance. Onthe figure is also shown the well-known A-weighting function and it canbe seen that apart from the total noise power of kT/C (which is given bythe microphone capacitance) also the location of the noise has animportance. That is, if the bias resistor can be made very large thenthe noise power inside the A-weighting function can be made very small.So the trick is to use a very large bias resistor to shape to appear asfar away from the centre of the A-weighing curve as possible.

Secondly, the PMOS generated noise is explained. From the PMOS device ofthe source follower originates a 1/f noise component and a white noisecomponent. These noise components can be made small in magnitude byincreasing the gate area of the device contributing with the noise. Bymaking the device very large both the 1/f and the white noise can bereduced. The 1/f noise can even be reduced to be completely negligibleby using a transistor with a large area. The consequence of making thetransistor very large is though that the signal is damped as themicrophone is capacitively loaded by the larger intrinsic capacitancecaused by the larger area. So the transistor should be large but not toolarge. Hence, an optimum exists for both 1/f noise and white noise.

This example shows the noise from a source follower stage but exactlythe same argumentation and tradeoffs apply for any CMOS preamplifiercoupled to a capacitive signal source.

FIG. 4 shows plots of the transfer function for open loop amplifiergain, feedback filter gain, loop gain and preamplifier gain. The plotsrefer to a circuit configuration shown in FIG. 1, especially thefeedback preamplifier comprising the amplifier 102 provided with thefeedback filter 103 to establish a feedback loop. The open loopamplifier gain is illustrated by the curve designated #1, feedbackfilter gain is illustrated by the curve designated #2, the loop gain isillustrated by the curve designated #3 and the overall preamplifier gainis illustrated by the curve designated #4.

The plots serves to illustrate the principle of a preamplifier with afeedback filter from a gain-frequency domain point of view. The plotsare shown for a first order filter configuration. Corresponding plotscan be obtained for higher order filter configurations.

The open-loop amplifier gain, curve #1, is the well-known dominatingpole type with a single dominating pole located at frequency F1 and witha very large gain at low frequencies.

The feedback filter characteristic, curve #2, comprises in its simplestfirst order form one pole and one zero. The pole being located at afrequency, F2, lower than the frequency, F3, of the zero. Basically, thefilter has a lower frequency region and a higher frequency region with alower gain than the low frequency region. Thus, the feedback filtercharacteristic shows a relatively high gain level below F2 and thusbelow a transition frequency range between F2 and F3 and a relativelylow gain level above the transition frequency range i.e. above F3.

In practical implementations the preamplifier has a transfer function,in the frequency domain, with a zero and a pole cf. curve #2; whereinthe pole is located in the range 0.1 Hz to 50 Hz or alternatively 0.1 Hzto 100 Hz or alternatively 0.1 to 200 Hz.

The combination of the open-loop gain and the feedback filtercharacteristic provides the loop-gain characteristic, curve #3. It canbe seen that the loop-gain is very large below frequency F2.

The overall gain transfer function of the amplifier in a feedbackconfiguration is shown by curve #4. Where the feedback configuration isestablished by means the feedback filter with a gain characteristic asshown in curve #2.

FIG. 5 a shows a first feedback filter. The feedback filter 103 forms afeedback circuit with an input port designated ‘a’ and an output portdesignated ‘b’. The input port, a, is connected to a ground referencevia a series connection of a first resistor R2, 502; a capacitor C, 503;and a second resistor R3, 501. The output port, b, is coupled to thecircuit node formed by the interconnection of the first resistor R, 502and the capacitor C, 503.

The feedback filter can be implemented in many ways but not all of themare equally easy to integrate on a chip. Especially filter types withseries resistors are difficult to implement as the component valuesneeded are difficultly implemented on a chip or semiconductor substrate.

The desired filter transfer function is a high-pass filter function.This is typically implemented using two resistors in series with acapacitor (see FIG. 5 a). At lower frequencies the transfer functionfrom port a to port b is close to one and at higher frequencies it isdetermined by the ratio of R2 and R3. In order to obtain low noise theresistors will have to be in the kOhm range and thus requiring thecapacitor value to be nF range to realize a desired cut-off frequency.Capacitors in the nF range would require excessive chip area, and suchsolutions are thus deemed not possible for a chip implementation.

FIG. 5 b shows a second feed-back filter for IC implementation. Thefeedback filter 103 forms a feedback circuit with an input portdesignated ‘a’ and an output port designated ‘b’. The filter has aconfiguration with an input port, a, connected to a series connection ofa first resistor R2, 507 and a second resistor R3, 508 which forms aresistor node at their interconnection. The input port is also connectedto a series connection of a first capacitor C1, 506 and second capacitorC2, 504 which forms a capacitor node at their interconnection. Thecapacitor node forms the output port. Additionally, the resistor nodeand capacitor node are interconnected by a resistor R1, 505.

For this configuration of the feed-back filter, the low frequencytransfer function from port a to port b is determined by the tworesistors R2, 507 and R3, 508. The high frequency transfer function isdetermined by C1, 506 and C2, 504. The cut-off frequency of the filtercan be set by R1, 505. If R1, 505 is chosen very large, the noise of thefilter will moved to very low frequencies and the audio band noise canthus be minimized without using excessively large capacitor values.Suitable ranges for implementation on a semiconductor substrate areC1=1-500 pF, C2=1-500 pF and R1=GOhm−Tohm.

FIG. 5 c shows a fourth feed-back filter for IC implementation. Thefeedback filter 103 forms a feedback circuit with an input portdesignated ‘a’ and an output port designated ‘b’. The filter has aconfiguration with an input port, a, connected to a series connection ofa first resistor R2, 507 and a second resistor R3, 508 which forms aresistor node at their interconnection. The input port is also connectedto a series connection of a first capacitor C1, 506 and second capacitorC2, 504 which forms a capacitor node at their interconnection. Thecapacitor node forms the output port. Additionally, the resistor nodeand capacitor node are interconnected by an active device 516 whichprovides ohmic impedance across a two-port circuit. The two-port circuitcomprises the active CMOS devices, or other type of active devices, 516and 517, and a current source 518. The active devices comprise arespective gate terminal, a source terminal and a drain terminal. Thegate terminals are interconnected at a node connected to the currentsource 518 and the drain terminal of the first device 517. The sourceterminals of the devices are interconnected, to provide the seconddevice 516 in a state where an ohmic resistance is provided between itsdrain and source terminal. This drain and source terminal constitutesthe two-port ohmic impedance. Cf. the description above, the cut-offfrequency of the filter 103 is set by the ohmic impedance of thetwo-port circuit 516, 516 and 518.

With respect to the two-port circuit, a MOS transistor in the trioderegion is biased to have impedance in the G ohms region is a goodsolution. It can be fairly well controlled and easily implemented in anyCMOS technology.

In our case a NMOS device will prove most sufficient because of the DCoperating levels. But in other case also PMOS devices can be used andeven symmetrical devices made by a combination of NMOS and PMOS devicescan be used. But it depends on the DC levels if completely symmetricaldevice can be used.

The only disadvantage of the non-symmetrical NMOS resistor is that it isnonlinear thus it will generate a low frequency signal (i.e. a dcoffset) when exposed to a large sinus. The amplitude of the lowfrequency signal generated by the NMOS resistor will be highly dependenton how the feedback filter is constructed. I.e. if the feedback factorof the capacitive feedback and the feedback factor of the resistivefeedback are equal then the signal across the NMOS resistor will be zeroand thus the generated low frequency signal will also be zero. But asargued earlier a large DC gain of the preamplifier will be a problem aslarge low frequency signals will be present at the input of thepreamplifier and thus overload the preamplifier. A DC feedback factorgiving a DC gain of two will effectively reduce the signal across theNMOS resistor by a factor of two and thus reduce the low frequencygenerated signal by at least a factor of two. However, DC gains in therange of one to five (1-5) will typically be suitable. That is, signalsgenerated due to non-linear signal processing depends typically on thesquare of the generating signal or on the cubic of the generatingsignal. So halving the signal across the NMOS resistor will reduce thegenerated signal by a factor of four.

Thereby a way of implementing a resistor in the Giga-ohms range whichcan be relatively well controlled is provided. Ordinary use of diodes,diode coupled MOS transistors etc. can implement extremely largeresistor values (in the order of Tera-ohms). However, these values arein fact too large for the purpose related to the IC implementedpreamplifier.

FIG. 5 d shows a third feed-back filter for IC implementation. In thisembodiment, a DC offset has been build into the feedback filter 103.Thereby it is possible to compensate for undesired DC levels that arisedue to leak currents. The DC offset is implemented by replacing theresistor R3 with a current source 513.

The total noise power from the filter 103 at the output port b (and thusat the inverting input of the amplifier 102) can be calculated to beKT/C where C is the total capacitance, K is Boltzman's constant and T isthe temperature in Kelvin.

The noise can as such not be minimized without increasing the capacitors(C1 and C2) and this has the consequence that the area of the chip willbe increased. This is not a feasible solution when a low cost solutionis needed. However, the noise power can be moved or shaped to appear atlower frequencies by increasing the resistor value R1. In this way thenoise from the resistive feedback network is filtered and the noise fromthe capacitors C1 and C2 (in fact the noise comes from R1) will befiltered and located at such low frequencies that the normally usedA-weighting function will suppress the noise. So this solution has bothlow area and low noise.

The resistor R1 should be large in order to get low noise for thesmallest possible area. But if the resistor is too large, the amplifierwill settle to slowly after a power on or after an overload. This willsignificantly degrade performance of the amplifier for a longer periodof time.

FIG. 6 shows a detailed view of the amplifier. The amplifier input stage601 comprises a differential pair of PMOS devices 603, 606. Thisdifferential pair will have to be optimized in both width and length asan optimum for 1/f noise and white noise exists (see FIG. 3). If needed,an offset can be built into the differential pair by adjusting theaspect ratio of the two transistors in the differential pair (see FIG.10 d). Alternatively or additionally, the mirroring factor of thecurrent mirror 604, 605 in the bottom can be adjusted. If the ratiobetween the aspect ratio of the differential pair transistors are A andthe current mirror factor is B, the offset of the amplifier will ben*Vt*In(A*B).

Various implementations of a differential input stage exist—forinstance, the NMOS current mirror 604, 605 can be replaced by aso-called folded cascode in combination with a PMOS current mirror.

At the output stage 602 of the amplifier, an output transistor 608 isconnected to the high impedent gain node. The function of this is to adgain and to isolate the high impedant node from the outside. Notice thatthe only device which has a varying current is the output transistor.Thereby the other transistors are biased by constant current sources.

Thus, a principle amplifier with a differential input stage and anoutput stage is described.

FIG. 7 a shows an amplifier with a feed-back filter and an inputclamping circuit. The input clamping circuit 701 is connected to theinput of the amplifier receiving the microphone signal (in this case thenon-inverting input). Which clamping circuit to use depends onavailability of technology and demands.

FIG. 7 b shows a diode based input clamping circuit. This circuit iscommonly known and has proven to work well. It consists of two crosscoupled diodes. The impedance/resistance around zero biasing is veryhigh and at typically 400 mV-600 mV the device starts to clamp thesignal—i.e. the impedance drops dramatically. The impedance is very highat zero biasing and it clamps the signal for large signal levels as itshould. The impedance may however be too large for some circuits(examples of up to 100 Tohms has been measured) and the clamping of thesignal happens very often at too low signal levels.

In that case the two following solutions might prove better. It ispossible to make new symmetrical high impedant devices by combining anyof the three implementations.

FIG. 7 c shows a PMOS based input clamping circuit. This implementationis basically an implementation of two cross-coupled diodes using MOSdevices 704, 705 instead.

FIG. 7 d shows a NPN transistor based input clamping circuit. Thisimplementation is basically an implementation of two cross-coupleddiodes using bipolar devices 706, 707 instead.

FIG. 8 a shows an amplifier with a feed-back filter and an output stage.The output stage 802 is part of the complete preamplifier and isembedded in a feedback loop comprising amplifier 801, the output stage802 and the feedback filter 103. The purpose of the output stage 802 isto isolate the internal circuit nodes of the preamplifier from the loadcoupled to the output terminal V1/out.

FIG. 8 b shows a common source output stage. The first example is acommon source stage with a miller compensation capacitor 805 and aresistor 804 for right half-plane zero compensation. This stage has theadvantage that the output swing can be very large, the DC gain is largeand frequency compensation is very easy to implement. The disadvantageis that parameters varies much with the load and the output swing. Thatis, if the load is resistive then the current will vary both with loadand the output swing. This has the consequence that the millercompensation will have to be designed for worst case conditions andparameters such as distortion PSR etc. will vary much with load andsignal swing.

FIG. 8 c shows a source follower output stage. The second example is asource follower stage based on an active device 806. This effectivelyisolates the internal circuit nodes from the load and the all theparameters are very stable both with varying load and output swing. Thedisadvantage with the source follower stage is that frequencycompensation requires large capacitors and that the output swing islimited compared to the common source stage.

FIG. 8 d shows an output stage with a combined common source and sourcefollower configuration. The third example is the combination of thecommon source stage and the source follower. This enables both easyfrequency compensation and stable performance. The only disadvantage isthe limited output swing compared to the simple common source stage.

FIG. 8 e shows a cascaded common source output stage. The fourth exampleis two common source stages in series, where the active device of thefirst stage is biased by a current source 810. In this case so-callednested miller compensation has to be used. This solution has more stableperformance than the simple common source stage but not as stable as thecombination of a common source stage and a source follower. But theoutput swing is just as good as the simple common source stage.

FIG. 9 shows the preamplifier with RF filters implemented. The RadioFrequency, RF, filters 901 comprise a capacitor 903 and a resistor 902coupled to receive an input signal at their interconnection at thecircuit node designated port ‘h’ and to provide an output signal at port‘i’. A ground reference is provided at port ‘j’. Thereby a first orderlow-pass filter diminishing RF signals is provided. However, otherhigher order filters can be used—e.g. 2^(nd), 3^(rd), and 4^(th) orderfilters.

With today's widespread use of mobile phones, microphones are exposedfor very power-full high frequency signals e.g. the RF GSM signal of amobile phone. And especially a microphone for a mobile is exposed tovery large RF signals as it is located very close to the antenna. It iswell know that nonlinearities in semiconductors can inter-modulate lowfrequency variation of RF signals into the low frequency bandwidth. Toexemplify this, a GSM phone is transmitting at periods of 217 Hz. If byexample a diode is exposed to a GSM signal then the nonlinearity of thediode together with the GSM signal will create a very power-full 217 Hzcomponent and harmonics of this. One of the most efficient ways ofreducing this problem is simply to prevent the GSM signal from reachingthe nonlinear semiconductor components. This can be implemented byadding RF filters 901 on every connection pad for the amplifier ASIC.

This approach is very effective but the problem is that these filtersbeside filtering the RF noise, also affect the performance of theamplifier. That is, output impedances increase, noise levels increaseetc. But in the case of a feedback amplifier this problem is greatlyreduced as the overall performance of the amplifier is greatlydetermined by the feedback filter and by the input stage. So if the highfrequency components can be prevented from accessing the input stage,the feedback loop will by itself suppress low frequency inter-modulatedsignals introduced elsewhere in the amplifier. Due to, typically, a highopen loop gain, the amplifier will have a performance which isunaffected even though very effective RF filters are added to the outputpads/connections.

Also on the input side, RF filters can be added, but here the noise ofthese are not suppressed by the loop gain. But as the overall amplifierstructure has lower noise, this can be traded off for a more effectiveRF filter on the input.

FIG. 10 a shows a preamplifier and feedback filter with DC offsetsimplemented. The purpose of the DC offsets 1001, 1002 and 1003 is to setthe DC bias voltage of the output of the amplifier. A DC offsetimplemented in the amplifier will be amplified by the DC closed loopgain of the amplifier thus setting the output DC level of the amplifier.In order to handle low-frequency signals and external offsets on theinput the DC offset of the amplifier will have to be designed for afairly large value. Which can be a problem when optimizing for lowestpossible noise. And also when optimizing for lowest possible chiparea—i.e. lowest cost.

More optimal is it to implement the offset in or just after theFB-filter. In this way the offset will not be multiplied by the DCclosed loop gain of the amplifier and also the amplifier can be designedfor lower noise without increasing the chip area. It can though be aproblem to implement all off the offsets in the filter so a combinationof an offset in the amplifier and an offset in the feedback filter willnormally prove to be optimal.

FIG. 10 b shows an amplifier configuration with DC level compensation atthe input terminal of the amplifier. A current source 1006 provides a DCcurrent through a resistor R5, 1004. The DC voltage level provided overthe resistor R5, 1004 will shift the DC level at the input of theamplifier 102.

There are basically two ways of implementing an offset at the input ofthe preamplifier. The first one is provided by a reference voltagesource filtered by a large resistor. The resistor can be implemented inmany ways. I.e. as diodes active devices etc.

The second one is offset implemented in the input differential pair.This can be done in many ways e.g. with a deliberate mismatch incurrents or sizes of in the input differential pairs as sketched in FIG.10 d. Alternatively, the DC offset can be provided by a device with a DCoffset in series with one of the sources.

FIG. 10 c shows a circuit which implements a high-ohmic resistor. Apartfrom diodes transistors etc. also active devices can be used as highresistive devices. A NMOS device 1009 biased in weak inversion will havean ohmic resistance between drain and source at port x1 and y1,respectively, approximately equal to A×nVt/Id, where a is the ratiobetween the two transistor aspect ratios. Id is the current in the biastransistor, nVt=39 mV for most CMOS processes. The NMOS device 1009 isbiased in weak inversion by means of a NMOS device 1008 biased by acurrent source 1009.

Thus, the active device 1009 is forced to be in a state, where a highohmic impedance is provided between its source and drain terminal. Anohmic impedance larger than 50 MOhm or larger than 100 MOhm or largerthan 500 MOhm is provided. Alternative embodiments using poly-diodeconfigurations are generally not preferred due to its specificimplementation semiconductor technology.

In this way Giga-ohm resistors can be implemented using CMOS devices.The disadvantage of this device is that it is asymmetrical. This can besomewhat compensated adding a symmetrical device across the NMOSresistor.

FIG. 10 d shows a sketch of an input stage of a differential amplifierconfiguration with DC level compensation at the input stage. The DClevel compensation is obtained by providing a difference between thecurrents I1 and I2 flowing through the differential pair comprised bytransistors 1011 and 1010, respectively. The current source 1012illustrates biasing of the differential pair.

FIG. 11 shows the amplifier in a configuration with a voltage pump. Avoltage pump Vpmp, 1101 is typically needed when there is no chargedelectret layer in the microphone as in e.g. a silicon microphone. Thevoltage pump Vpmp, 1101 provides a pumped voltage as a bias voltage Vbvia the resistor Rc, 1102. A capacitor Cc operates together with theresistor Rc to decouple noise.

When there is no electret layer in the microphone, an external bias isneeded and can be supplied by a voltage pump integrated on the samesemiconductor substrate as the preamplifier. Voltage pumps are normallyquite noisy and thus a decoupling filter is needed. This filter canconsist of a decoupling capacitor Cc and a large resistor, Rc. Todecouple the noise of the voltage pump 1101, a filter with a very lowcut-off frequency is needed. And thus it settles very slowly duringpower up. That is, a very large low frequent signal will be present onthe input of the amplifier for a substantial period of time. So, thepreamplifier with a low gain at low frequencies again proves to be verybeneficial.

Some microphone types requires a bias voltage in order to function e.g.silicon microphones. Such as bias voltage is normally higher than thesupply voltage. In fact in can be as high as 30V and thus many timeshigher than the power supply. Such a bias voltage is generated using avoltage pump which typically is comprised by a Dickson pump and anoscillator to provide a clock signal to the Dickson pump. However, theclock signal can be provided an external oscillator—in which case aseparate input terminal of the semiconductor substrate typically isneeded.

If the microphone is biased at a high DC voltage, a DC couplingcapacitance is needed between the amplifier and the microphone as thenamplifier in nearly all cases are not able to handle the large DC levelwithout overload. Furthermore by integrating everything on the same chipthe total performance can be optimized giving the best possibleperformance.

FIG. 12 shows a microphone comprising an electret microphone member anda differential amplifier. The electret microphone member is biased via abias resistor 104 that is coupled to a bias voltage Vb. Thereby anelectric charge is provided to the membrane or movable member of themicrophone 105, Cmic. A signal provided in response to a sound pressureon the microphone and thus making the membrane move is provided to anamplifier 1201. The amplifier 101 is characterised by having a gaincharacteristic with relative low gain for frequencies below an audiblerange and a relative high gain for frequencies in the audible range.Preferably, the gain characteristic descents as a 1^(st), 2^(nd),3^(rd), 4^(th), or higher order below the audible range. In additionthereto the amplifier is characterised by processing a low frequencymicrophone signal as a common-mode signal and a high frequencymicrophone signal as a differential mode signal. Thereby low frequencycomponents are effectively suppressed. The differential output signal atterminals φ and φ* is provided as a microphone preamplifier outputsignal.

FIG. 13 a shows a differential amplifier with input and output terminalsand signals illustrating a low frequency behaviour of the differentialamplifier. The signal processing of amplifier 101 at low frequencies isillustrated. The curve 201 illustrates a microphone signal in thetime-domain input to the amplifier (φ) and at a low frequency. Thecurves 1302 and 1303 illustrates that respective outputs (φ, φ*) of theamplifier are substantial in phase and thus represent a common-modedifferential signal.

FIG. 13 b shows the differential amplifier with input and outputterminals and signals illustrating a high frequency behaviour of thedifferential amplifier. The signal processing of amplifier 1201 at high,audio band, frequencies is illustrated. The curve 1301 illustrates amicrophone signal in the time-domain input to the amplifier (φ) and at aaudio frequency. The curves 1302 and 1303 illustrates that respectiveoutputs (φ, φ*) of the amplifier are substantial 180 degrees out ofphase and thus represent a differential-mode differential signal.

FIG. 14 shows a portion of a digital microphone comprising an electretmicrophone member and a differential amplifier in a first configuration.The microphone Cmic, 105 provides a signal to the differential amplifier1408 via a DC blocking capacitor 1404. The differential amplifier iscoupled as an instrumental amplifier and comprises a first amplifier1401 and a second amplifier 1402. A feedback circuit comprisingcomponents 1405, 1406 and 1407 implements a feedback filter with acharacteristic as disclosed in the above. Additionally, a phase-shiftercircuit PD(f), 1403 implements a frequency dependent phase delay toestablish the signal processing explained in connection with FIG. 13 aand FIG. 13 b.

The two-terminal output (φ, φ*) of the differential amplifier isprovided to a difference amplifier 1409 which by means of resistors1410, 1411 and 1412 and operational amplifier 1413 provides asingle-ended output signal Vo with respect to the ground reference, Gnd.

The amplifiers 1408 and 1409 can be implemented on a commonsemiconductor substrate, but the differential signal is preferablyrouted where the electrical environment is noisy e.g. from the chipcarrying the preamplifier to another chip for further signal processing.At this other chip the differential signal can be converted to asingle-ended signal by means of the amplifier 1409.

FIG. 15 shows a portion of a digital microphone comprising an electretmicrophone member and a differential amplifier in a secondconfiguration. The electret microphone member is biased via a biasresistor 104 that is coupled to a bias voltage Vb. Thereby an electriccharge is provided to the membrane or movable member of the microphoneCmic, 105. In order to block the DC bias voltage, Vb, from the input ofthe differential amplifier 1510 a capacitor 1404 is applied. Thedifferential amplifier 1510 is configured as a so-called instrumentationamplifier wherein two operational amplifiers 1501 and 1502 each arecoupled with a feedback path from their respective outputs φ, φ* totheir respective inverting input terminal. The inverting inputs (−) ofthe operational amplifiers are coupled together by means of a capacitor1505. A non-inverting input of the one operational amplifier 1501 iscoupled to receive the microphone signal via the DC-blocking capacitor1404. A non-inverting input of the other operational amplifier 1502 iscoupled to receive a feedback signal from the output φ of the otheroperational amplifier via a resistor 1508. The non-inverting input isalso coupled to ground by means of a capacitor 1509.

The feedback path of the operational amplifier 1501 comprises a resistor1503 and a capacitor 1504 coupled in parallel to constitute a firstorder filter. Likewise, the feedback path of the operational amplifier1502 comprises a resistor 1507 and a capacitor 1506 coupled in parallelto constitute a first order filter.

The RC network comprising resistor 1508 and capacitor 1509 is configuredto provide a frequency dependent phase shifting of the signal.

The phase shift between the one side, constituted around operationalamplifier 1501, of the differential amplifier and the other side,constituted around operational amplifier 1502 is implemented partly bycapacitor 1505 and partly by the RC filter 1508, 1509. Thus, the phaseshift is obtained by a phase shifter, capacitor 1505, coupled betweeninputs of the differential amplifier and a phase shifter, capacitor 1509and resistor 1508, cross coupled between an output of one side of thedifferential amplifier and an input of the opposite side of thedifferential amplifier. Thus, the effective phase shift is obtained bymeans of two phase shifters. However, one of such two coupled phaseshifters may be sufficient to establish the effective phase shift.Likewise, other configurations of phase shifters can be embodied withoutdeparting from the scope of the invention.

FIG. 16 shows a portion of a digital microphone comprising an electretmicrophone member and a differential amplifier with a feedback filter.In this illustration a differential amplifier 1607 with a first and asecond operational amplifier is shown with a filter block 1603. Thefilter block 1603 implements feedback paths of the respectiveoperational amplifiers and coupling of the inverting inputs of therespective operational amplifiers 1601 and 1602. The filter blockcomprises input ports m, n and output ports k, l.

The filter block can implement a filter with two feedback paths of anyorder e.g. a 1^(st) order, 2^(nd) order, 3^(rd) order, 4^(th) order orany higher order.

FIG. 17 shows a preferred embodiment of a feedback filter. The feedbackfilter can implement the filter block 1603 of FIG. 16. The feedbackfilter comprises a first path with resistor 1701 coupled in parallelwith a capacitor 1702 and a second path with resistor 1704 coupled inparallel with a capacitor 1703. The first path extends between port mand k and the second path extends between port n and l.

FIG. 18 is a schematic view of a microphone with an integrated circuitand a microphone member. The integrated circuit 1802 comprises thepreamplifier as disclosed above and is embodied on a semiconductorsubstrate or a chip.

FIG. 19 is a schematic view of a microphone with an integrated circuitand a MEMS microphone member. The microphone 1902 comprises a MEMSmicrophone member 1903 integrated on a first substrate and thepreamplifier circuitry 1901 integrated on a second substrate. Thepreamplifier circuitry comprises one of the different embodimentsdisclosed above i.e. comprising a preamplifier with a feedback circuitand .e.g. a voltage pump and/or a feedback circuit, where thepreamplifier is a differential amplifier or a single-ended amplifier.

It should be noted that the MEMS microphone member 1903 and themicrophone preamplifier 1901 can be integrated on a single semiconductorsubstrate.

Generally, it should be noted that embodiments of the invention maycomprise one or more of the described features. For instance, thepreamplifier may comprise one or more of the following features:

-   -   A DC offset in the preamplifier stage;    -   A DC offset in the feedback filter;    -   A DC offset in both the preamplifier stage and the feedback        filter;    -   A voltage pump;    -   A voltage pump in combination with a DC offset;    -   A radio frequency, RF, filter coupled to the following circuit        nodes:        -   a non-inverting amplifier input; and/or        -   an inverting amplifier input; and/or        -   a filter input; and/or        -   an amplifier output.    -   An input bias element.

It should be noted that the invention is not limited to the disclosedembodiments.

The above features may be applied in embodiments of a preamplifierconfiguration that comprises a gain stage with a feedback filter, wherethe configuration has a relatively low gain response for frequenciesbelow an audio band and has a relatively high and substantially flatgain response in the audio band. The audio band can be defined to be anyband within the typical definition of an audio band. A typicaldefinition can be 20 Hz to 20 KHz. Exemplary lower cut-off frequenciesfor an audio band can be: 20 Hz, 50 Hz, 80 Hz, 100 Hz, 150 Hz, 200 Hz,250 hz. Exemplary upper cutoff frequencies the an audio band could be 3KHz, 5 KHz, 8 KHz, 10 KHz, 18 KHz, 20 KHz. By substantial flat is meantgain response variations within approximately +/−1 dB; +/−3 dB; +/−4 dB;+/−6 dB. However, other additional values of variation can be used todefine the term ‘substantial flat’.

In the above different preamplifier configurations have been disclosed.These configurations comprise different input/output terminalconfigurations e.g. a two-terminal configuration. However, it should benoted that three, four or more terminals can be provided forinput/output of signals to microphone and preamplifier. Especially, itshould be noted that separate terminals can be provided for supplyvoltage (at a first terminal) and preamplifier output (at a secondterminal). In case of a differential preamplifier output two terminalsfor the output signals can be provided in addition to a terminal forpower supply. A separate terminal is provided for a ground reference.This ground reference is typically, but not always, shared by the powersupply and output signal.

1. A microphone preamplifier, comprising: a semiconductor substrate; adifferential amplifier integrated on the semiconductor substrate havingan input stage with an inverting input terminal and a non-invertinginput terminal and an output stage with an output terminal wherein anaudio frequency input signal from a microphone is to be applied to thenon-inverting input terminal; a feedback circuit also integrated on thesemiconductor substrate comprising an active device which provides anohmic impedance across a two-port circuit having a low-pass frequencytransfer function that couples a part of the input signal from theoutput terminal back to the inverting input terminal; and a DC offsetimplemented in at least one of at the input stage, in the input stage,or in the feedback circuit to set the DC bias voltage at the outputterminal.
 2. A microphone preamplifier according to claim 1, wherein thefeedback circuit is a filter with a transfer function, in the frequencydomain, with a zero and a pole; wherein the zero is located at a higherfrequency than the pole.
 3. A microphone preamplifier according to claim1, wherein the preamplifier has a transfer function, in the frequencydomain, with a zero and a pole; wherein the pole is located in the range0.1 Hz to 50 Hz or 0.1 Hz to 100 Hz or 0.1 to 200 Hz.
 4. A microphonepreamplifier according to claim 1, wherein the feedback circuit is afilter which, in the frequency domain, has a relatively high gain levelbelow a transition frequency range and a relatively low gain level abovethe transition frequency range.
 5. A microphone preamplifier accordingto claim 4, wherein the transition frequency range is located below afrequency of about 100 Hz.
 6. A microphone preamplifier according toclaim 4, wherein the transition frequency range is located below afrequency of 40 Hz.
 7. A microphone preamplifier according to claim 1,wherein the feedback circuit is an active filter.
 8. A microphonepreamplifier according to claim 1, wherein the feedback circuit is apassive filter.
 9. A microphone preamplifier according to claim 1,wherein the feedback circuit comprises a configuration with a first anda second active device and a current source, where the devices comprisea respective gate terminal, a source terminal and a drain terminal, andwhere the gate terminals are interconnected at a node connected to thecurrent source and the drain terminal of the first device, and where thesource terminals are interconnected, to provide the second device in astate where an ohmic resistance is provided between its drain and sourceterminal.
 10. A microphone preamplifier according to claim 1, whereinthe feedback circuit comprises a filter with an input port connected toa series connection of a first and second resistor which forms aresistor node at their interconnection, and connected to a seriesconnection of a first and second capacitor which forms a capacitor nodeat their interconnection; and an output port at the capacitor node;wherein the resistor node and capacitor node are interconnected by anactive device which provides an ohmic impedance across a two-portcircuit.
 11. A microphone preamplifier according to claim 1, wherein thefeedback circuit comprises a source providing a DC offset.
 12. Amicrophone preamplifier according to claim 1, wherein the feedbackcircuit comprises a filter with a source that provides a DC offset. 13.A microphone preamplifier according to claim 1, wherein the DC offset isprovided at the inverting input terminal by a circuit configurationcomprising a current source coupled, at the circuit node of theinverting input terminal a resistor, a diode, or an active device whichprovides an ohmic impedance across a two-port circuit.
 14. A microphonepreamplifier according to claim 13, wherein the active deviceconstitutes a second device in a configuration with a first and thesecond active device and a current source, where the devices comprise arespective gate terminal, a source terminal and a drain terminal, andwhere the gate terminals are interconnected at a node connected to thecurrent source and the drain terminal of the first device, and where thesource terminals are interconnected, to provide the second device in astate where an ohmic resistance is provided between its drain and sourceterminal.
 15. A microphone preamplifier according to claim 1, whereinthe input stage comprises a first and second current path for therespective signal inputs, and wherein a DC offset is provided byestablishing different DC currents through the first and second currentpath of the input stage.
 16. A microphone preamplifier according toclaim 1, wherein the differential amplifier is configured to convert aninput signal into a common mode signal for low frequencies and into adifferential signal for audio frequencies.
 17. A microphone preamplifieraccording to claim 1, wherein the differential amplifier is configuredas an instrumentation type amplifier with two inputs and a first and asecond output, wherein the first and second input is arranged to receivea microphone signal, but wherein the inputs are coupled to receive themicrophone signals substantially in phase at relatively low frequenciesand substantially out of phase at relatively high frequencies.
 18. Amicrophone preamplifier according to claim 1, wherein the differentialamplifier is configured to provide frequencies below an audio band ascommon mode signals and audio band signals as differential mode signals.19. A microphone preamplifier according to claim 1, wherein a phaseshifter is coupled between the inputs of the differential amplifier. 20.A microphone preamplifier according to claim 1, wherein a phase shifteris cross coupled between an output of one side of the differentialamplifier and an input of the opposite side of the differentialamplifier.
 21. A microphone preamplifier according to claim 1, furthercomprising a voltage pump integrated on the semiconductor substrate. 22.A microphone module comprising the microphone preamplifier according toclaim 1, and further comprising an electret microphone configured toprovide a microphone signal, responsive to a sound pressure on theelectret microphone, to the microphone preamplifier.
 23. A microphonemodule comprising the microphone preamplifier according to claim 1, andfurther comprising an electret microphone mounted inside a space formedby a cartridge, and wherein the microphone preamplifier is integratedwithin the microphone module.
 24. A microphone preamplifier according toclaim 1, and further comprising a MEMS microphone member to provide amicrophone signal, responsive to a sound pressure on the MEMSmicrophone, to the microphone preamplifier.
 25. A microphonepreamplifier according to claim 24, wherein the MEMS microphone memberand the microphone preamplifier are integrated on a semiconductorsubstrate.
 26. The microphone preamplifier as claimed in claim 1 whereinsaid differential amplifier has a high pass frequency transfer functionand said feedback circuit has a low-pass frequency transfer functionthat reduces the low frequency output of the preamplifier.
 27. Amicrophone preamplifier according to claim 1, wherein the DC offset isimplemented by a combination of DC offsets selected from a group of atthe input stage, in the input stage and in the feedback circuit.
 28. Amicrophone preamplifier according to claim 1, wherein the differentialamplifier is configured to convert an input signal into a common modesignal for low frequencies and into a differential signal for audiofrequencies.
 29. A microphone preamplifier according to claim 1, andfurther comprising a phase shifter cross coupled between an output ofone side of the differential amplifier and an input of the opposite sideof the differential amplifier.
 30. A microphone preamplifier comprising:a semiconductor substrate; a differential amplifier integrated on thesemiconductor substrate having an input stage with a first signal inputterminal and a second signal input terminal and an output stage with anoutput terminal wherein an audio frequency input signal from amicrophone is to be applied to the second signal input terminal; afeedback circuit also integrated on the semiconductor substratecomprising an active device which provides an ohmic impedance across atwo-port circuit having a low-pass frequency transfer function thatcouples a part of the input signal from the output terminal back to thefirst signal input terminal; and a DC offset implemented in at least oneof at the input stage, in the input stage, or in the feedback circuit toset the DC bias voltage at the output terminal, wherein the preamplifieris configured to receive the microphone signal via an input bias elementwhich has relatively high ohmic impedance when the microphone signal isrelatively small in magnitude and relatively low ohmic impedance whenthe microphone signal is relatively high in magnitude.
 31. A microphonepreamplifier according to claim 30, wherein the bias element isconfigured by two cross-coupled diodes.
 32. A microphone preamplifieraccording to claim 30, wherein the bias element is configured by twocross-coupled bipolar transistors.
 33. A microphone preamplifieraccording to claim 30, wherein the bias element is configured by twocross-coupled Metal Oxide Semiconductor, MOS, devices.
 34. A microphonepreamplifier according to claim 33, wherein a phase shifter is coupledbetween a signal node, substantially in phase with an input signal tothe amplifier, and an input terminal of an opposite side of thedifferential amplifier.
 35. A microphone preamplifier according to claim30, wherein the input stage comprises an inverting input and anon-inverting input, wherein the non-inverting input is the secondsignal input terminal and is to receive the microphone signal, and theinverting input is the first signal input terminal and is to receive afeedback signal provided by the feed-back circuit.
 36. A microphonepreamplifier according to claim 30, wherein the feedback circuit is afilter with a transfer function, in the frequency domain, with a zeroand a pole; wherein the zero is located at a higher frequency than thepole.
 37. A microphone preamplifier according to claim 30, wherein thepreamplifier has a transfer function, in the frequency domain, with azero and a pole; wherein the pole is located in the range 0.1 Hz to 50Hz or 0.1 Hz to 100 Hz or 0.1 to 200 Hz.
 38. A microphone preamplifieraccording to claim 30, wherein the feedback circuit is a filter which,in the frequency domain, has a relatively high gain level below atransition frequency range and a relatively low gain level above thetransition frequency range.
 39. A microphone preamplifier according toclaim 30, wherein the transition frequency range is located below afrequency of about 100 Hz.
 40. A microphone preamplifier according toclaim 30, wherein the transition frequency range is located below afrequency of 40 Hz.
 41. A microphone preamplifier according to claim 30,wherein the feedback circuit is an active filter.
 42. A microphonepreamplifier according to claim 30, wherein the feedback circuit is apassive filter.
 43. A microphone preamplifier according to claim 30,wherein the feedback circuit comprises a configuration with a first anda second active device and a current source, where the devices comprisea respective gate terminal, a source terminal and a drain terminal, andwhere the gate terminals are interconnected at a node connected to thecurrent source and the drain terminal of the first device, and where thesource terminals are interconnected, to provide the second device in astate where an ohmic resistance is provided between its drain and sourceterminals.
 44. A microphone preamplifier according to claim 30, whereinthe feedback circuit comprises a filter with an input port connected toa series connection of a first and second resistor which forms aresistor node at their interconnection, and connected to a seriesconnection of a first and second capacitor which forms a capacitor nodeat their interconnection; and an output port at the capacitor node;wherein the resistor node and capacitor node are interconnected by anactive device which provides an ohmic impedance across a two-portcircuit.
 45. A microphone preamplifier according to claim 30, whereinthe feedback circuit comprises a source providing a DC offset.
 46. Amicrophone preamplifier according to claim 30, wherein the feedbackcircuit comprises a filter with a source that provides a DC offset. 47.A microphone preamplifier according to claim 30, wherein the DC offsetis provided at the first signal input terminal by a circuitconfiguration comprising a current source coupled, at the circuit nodeof the first signal input terminal a resistor, a diode, or an activedevice which provides an ohmic impedance across a two-port circuit. 48.A microphone preamplifier according to claim 47, wherein the activedevice constitutes a second device in a configuration with a first andthe second active device and a current source, where the devicescomprise a respective gate terminal, a source terminal and a drainterminal, and where the gate terminals are interconnected at a nodeconnected to the current source and the drain terminal of the firstdevice, and where the source terminals are interconnected, to providethe second device in a state where an ohmic resistance is providedbetween its drain and source terminals.
 49. A microphone preamplifieraccording to claim 30, wherein the input stage comprises a first andsecond current path for the respective signal inputs, and wherein a DCoffset is provided by establishing different DC currents through thefirst and second current path of the input stage.
 50. A microphonepreamplifier according to claim 30, wherein the differential amplifieris configured as an instrumentation type amplifier with two inputs and afirst and a second output, wherein the first and second input isarranged to receive a microphone signal, but wherein the inputs arecoupled to receive the microphone signals substantially in phase atrelatively low frequencies and substantially out of phase at relativelyhigh frequencies.
 51. A microphone preamplifier according to claim 30,wherein the differential amplifier is configured to provide frequenciesbelow an audio band as common mode signals and audio band signals asdifferential mode signals.
 52. A microphone preamplifier according toclaim 30, and further comprising a phase shifter coupled between inputsof the differential amplifier.
 53. A microphone preamplifier accordingto claim 30, and further comprising a voltage pump integrated on thesemiconductor substrate.
 54. A microphone module comprising themicrophone preamplifier according to claim 30, and further comprising anelectret microphone configured to provide a microphone signal,responsive to a sound pressure on the electret microphone, to themicrophone preamplifier.
 55. A microphone module comprising themicrophone preamplifier according to claim 30, and further comprising anelectret microphone mounted inside a space formed by a cartridge, andwherein the microphone preamplifier is integrated within the microphonemodule.
 56. A microphone preamplifier according to claim 30, and furthercomprising a MEMS microphone member to provide a microphone signal,responsive to a sound pressure on the MEMS microphone, to the microphonepreamplifier.
 57. A microphone preamplifier according to claim 56,wherein the MEMS microphone member and the microphone preamplifier areintegrated on a semiconductor substrate.
 58. The microphone preamplifieras claimed in claim 30 wherein said differential amplifier has a highpass frequency transfer function and said feedback circuit a low-passfrequency transfer function that reduces the low frequency output of thepreamplifier.
 59. A microphone preamplifier according to claim 30,wherein the DC offset is implemented by a combination of DC offsetsselected from a group of at the input stage, in the input stage and inthe feedback circuit.